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10-Bit, 65/80/105 MSPS Dual A/D Converter
Preliminary Technical Data
FEATURES
Integrated Dual 10-Bit A-to-D Converters Single 3 V Supply Operation (2.7 V to 3.3 V) SNR = 58 dBc (to Nyquist, AD9216-105) SFDR = 75 dBc (to Nyquist, AD9216-105) Low Power: 280mW at 105MSPS Differential Input with 500 MHz 3 dB Bandwidth Exceptional Cross Talk Immunity > 75dB Flexible Analog Input: 1 V p-p to 2 V p-p Range Offset Binary or Twos Complement Data Format Clock Duty Cycle Stabilizer
AVDD VIN+_A VIN- _A REFT_A REFB_A VREF SENSE AGND REFT_B REFB_B VIN+_B VIN-_B SHA ADC + 0.5V AGND
AD9216
specified over the industrial temperature range (-40C to +85C).
SHA
ADC
Output Mux/ Buffers
10
10
OTR_A D9A-D0A OEB_A MUX_SELECT CLK_A CLK_B DCS SHARED_REF PWDN_A PWDN_B DFS OTR_B
Clock Duty Cycle Stabilizer Mode Control
Output Mux/ Buffers
APPLICATIONS
Ultrasound Equipment IF Sampling in Communications Receivers: 3G, Radio Point-to-Point, LMDS, MMDS Battery-Powered Instruments Hand-Held Scopemeters Low Cost Digital Oscilloscopes
AD9216
10
10
D10B -D0B OEB_B
DRVDD DRGND
Figure 1. Functional Block Diagram
GENERAL DESCRIPTION
The AD9216 is a dual, 3 V, 10-bit, 65/80/105 MSPS analog-todigital converter. It features dual high performance sample-and hold amplifiers and an integrated voltage reference. The AD9216 uses a multistage differential pipelined architecture with output error correction logic to provide 10-bit accuracy and guarantee no missing codes over the full operating temperature range at up to 105 MSPS data rates. The wide bandwidth, differential SHA allows for a variety of user selectable input ranges and offsets including single-ended applications. It is suitable for various applications including multiplexed systems that switch full-scale voltage levels in successive channels and for sampling inputs at frequencies well beyond the Nyquist rate. Dual single-ended clock inputs are used to control all internal conversion cycles. A duty cycle stabilizer is available on the AD9216 (all speed grades) and can compensate for wide variations in the clock duty cycle, allowing the converters to maintain excellent performance. The digital output data is presented in either straight binary or twos complement format. Out-of-range signals indicate an overflow condition, whic h can be used with the most significant bit to determine low or high overflow. Fabricated on an advanced CMOS process, the AD9216 is available in a space saving 64-lead LFCSP (9x9) and is
Rev. PrD_6/15/2004
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective companies.
PRODUCT HIGHLIGHTS
1. Pin compatible with AD9238, dual 12-bit 20/40/65MSPS ADC and AD9248, dual 14-bit 20/40/65MSPS ADC. Speed grade options off 105 MSPS, 80 MSPS, and 65 MSPS allow flexibility between power, cost, and performance to suit an application. Low power consumption: AD9216-105: 105 MSPS = 280 mW AD9216-80: 80 MSPS = 238 mW AD9216-65: 65 MSPS = 215mW 4. The patented SHA input maintains excellent performance for input frequencies up to 100 MHz and can be configured for single-ended or differential operation. Typical channel isolation of 75 dB @ fIN = 10 MHz. The clock duty cycle stabilizer maintains performance over a wide range of clock duty cycles.
2.
3.
5. 6.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 (c) 2004 Analog Devices, Inc. All rights reserved.
AD9216
TABLE OF CONTENTS General Description .............................................................. 1 Product Highlights ................................................................ 1 DC Specifications (Continued) ............................................. 4 Switching Specifications ....................................................... 4 AC Specifications ................................................................. 5 Absolute Maximum Ratings ..................................................... 7 ESD Caution ......................................................................... 7 Terminology........................................................................ 10 Typical Performance CharacteristiC PLOTS (TBD).............. 12 Equivalent Circuits ................................................................. 13
Preliminary Technical Data
Theory of Operation............................................................ 13 Analog Input ....................................................................... 13 Clock Input and Considerations .......................................... 15 Power Dissipation and Standby Mode ................................ 15 Digital Outputs.................................................................... 15 Timing ................................................................................ 16 Data Format ........................................................................ 16 Voltage Reference............................................................... 17 Evaluation Board Diagrams (TBD) ........................................ 19 Outline Dimensions ................................................................ 20
REVISION HISTORY
PrA: Initial Version PrB: included specification tables, ordering guide, package and pin configuration and Theory of operation sections. PrC: Corrected pin configuration figure (Fig3) pin naming errors , updated supply spec, corrected timing diagram and latency. PrD: Removed 120MSPS Grade, Updated DCS,OEB_B pin descriptions, updated input referred noise, Demux Timing Diagram needs updating
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Preliminary Technical Data AD9216-SPECIFICATIONS
DC SPECIFICATIONS
AD9216
Table 1. (AVDD = 3 V, DRVDD = 2.5 V, Maximum Sample Rate, CLK_A = CLK_B; AIN = -0.5 dBFS Differential Input, 1.0 V Internal Reference, TMIN to TMAX, unless otherwise noted.)
Parameter RESOLUTION ACCURACY No Missing Codes Guaranteed Offset Error 1 Gain Error 2 Differential Nonlinearity (DNL) Integral Nonlinearity (INL)
2
Temp Full Full Full Full Full 25C Full 25C Full Full Full Full Full Full 25C 25C Full Full Full Full
Test Level VI VI VI IV V I V I V V VI V V V V V IV IV V V
AD9216BCP-65/80 Min Typ Max 10 10 0.3 1.0 0.5 0.5 0.5 0.5 15 30 5 0.8 2.5 0.1 0.8 0.4 1 2 2 7 35 TBD TBD TBD TBD
AD9216BCP-105 Min Typ Max 10 10 0.30 1.0 0.5 0.5 0.5 0.5 15 30 5 0.8 2.5 0.1 0.8 0.4 1 2 2 7 35 TBD TBD TBD TBD
Unit Bits Bits % FSR % FSR LSB LSB LSB LSB ppm/C ppm/C mV mV mV mV LSB rms LSB rms V p-p V p-p pF k?
TEMPERATURE DRIFT Offset Error 1 Gain Error INTERNAL VOLTAGE REFERENCE Output Voltage Error (1 V Mode) Load Regulation @ 1.0 mA Output Voltage Error (0.5 V Mode) Load Regulation @ 0.5 mA INPUT REFERRED NOISE Input Span = 1 V Input Span = 2.0 V ANALOG INPUT Input Span = 1.0 V Input Span = 2.0 V 3 Input Capacitance REFERENCE INPUT RESISTANCE POWER SUPPLIES Supply Voltages AVDD DRVDD Supply Current 2 IAVDD 2 IDRVDD PSRR POWER CONSUMPTION 4 DC Input 2 Sine Wave Input 5 Standby Power MATCHING CHARACTERISTICS Offset Error Gain Error
1 2 3
Full Full Full Full Full Full Full Full Full Full
IV IV V V V V VI V V V
2.7 2.25
3.0 2.5 TBD/TBD TBD/TBD 0.01 TBD/TBD 215/238 1/1 0.1 0.05
3.3 3.6
2.7 2.25
3.0 2.5 TBD TBD 0.01 TBD 280 1 0.1 0.05
3.3 3.6
V V mA mA % FSR mW mW mW % FSR % FSR
Gain error and gain temperature coefficient are based on the A/D converter only (with a fixed 1.0 V external reference). Measured at maximum clock rate with a low frequency sine wave input and approximately 5 pF loading on each output bit. Input capacitance refers to the effective capacitance between one differential input pin and AVSS. Refer to Figure xx for the equivalent analog input structure. 4 Measured with dc input at maximum clock rate. 5 Standby power is measured with the CLK_A and CLK_B pins inactive (i.e., set to AVDD or AGND). Specifications subject to change without notice.
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AD9216
DC SPECIFICATIONS (CONTINUED)
Preliminary Technical Data
Table 2. (AVDD = 3 V, DRVDD = 2.5 V, Maximum Sample Rate, CLK_A = CLK_B; AIN = -0.5 dBFS Differential Input, 1.0 V Internal Reference, TMIN to TMAX, unless otherwise noted.)
Parame ter LOGIC INPUTS High Level Input Voltage Low Level Input Voltage High Level Input Current Low Level Input Current Input Capacitance 1 LOGIC OUTPUTS DRVDD = 2.5V High Level Output Voltage Low Level Output Voltage
1
Temp Full Full Full Full Full
Test Level IV IV IV IV IV
AD9216BCP-65/80 Min Typ Max 2.0 - 10 - 10 2 0.8 +10 +10
AD9216BCP-105 Min Typ Max 2.0 - 10 - 10 2 0.8 +10 +10
Unit
V V A A pF
Full Full
IV IV
2.45 0.05
2.45 0.05
V V
Output Voltage Levels measured with 5 pF load on each output. Specifications subject to change without notice.
SWITCHING SPECIFICATIONS
Table 3. Switching Specifications
Parameter SWITCHING PERFORMANCE Max Conversion Rate Min Conversion Rate CLK Period 1 CLK Pulsewidth High 1 CLK Pulsewidth Low DATA OUTPUT PARAMETER 2 Output Delay (tPD) Pipeline Delay (Latency) Aperture Delay (tA) Aperture Uncertainty (tJ) 3 Wake-Up Time OUT-OF-RANGE RECOVERY TIME
1 2
Temp Full Full Full Full Full Full Full Full Full Full Full
Test Level VI V V V V VI V V V V V
AD9216BCP-65/80 Min Typ Max 65/80 1 15.4/12.2 6.2/5 6.2/5 2.0 4.8 6 1.0 0.5 2.5 2 6.0
AD9216BCP-105 Min Typ Max 105 1 9.5 4.2 4.2 2.0 4.8 6 1.0 0.5 2.5 2 6.0
Unit MSPS MSPS ns ns ns ns Cycles ns ps rms ms
The AD9216 has a duty cycle stabilizer circuit that, when enabled, corrects for a wide range of duty cycles (see TPC xx). Output delay is measured from CLOCK 50% transition to DATA 50% transition, with a 5 pF load on each output. 3 Wake-up time is dependent on the value of the decoupling capacitors; typical values shown with 0.1 F and 10 F capacitors on REFT and REFB. Specifications subject to change without notice.
N N-1 ANALOG INPUT N+1 N+2 N+8 N+3 N+4 N+5 N+6 N+7
tA
CLK DATA OUT
N-8
N-7
N-6
N-5
N-4
N-3
N-2 t
PD
N-1
N
N+1
Figure 2. Timing Diagram
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Preliminary Technical Data
AC SPECIFICATIONS
AD9216
Table 4. (AVDD = 3 V, DRVDD = 2.5 V, Maximum Sample Rate, CLK_A = CLK_B; AIN = -0.5 dBFS Differential Input, 1.0 V Internal Reference, TMIN to TMAX, unless otherwise noted.)
Parameter SIGNAL-TO-NOISE RATIO fINPUT = 2.4 MHz fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz fINPUT = 100 MHz SIGNAL-TO-NOISE AND DISTORTION RATIO fINPUT = 2.4 MHz fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz fINPUT = 100 MHz EFFECTIVE NUMBER OF BITS (ENOB) fINPUT = 2.4 MHz fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz fINPUT = 100 MHz TOTAL HARMONIC DISTORTION fINPUT = 2.4 MHz fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz fINPUT = 100 MHz WORST HARMONIC (2nd or 3rd) fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz SPURIOUS FREE DYNAMIC RANGE fINPUT = 2.4 MHz fINPUT = 19.6 MHz fINPUT = 32.5 MHz fINPUT = 69 MHz Temp 25C Full 25C Full 25C Full 25C 25C 25C Full 25C Full 25C Full 25C 25C 25C Full 25C Full 25C Full 25C 25C 25C Full 25C Full 25C Full 25C 25C Full Full Full 25C Full 25C Full 25C Full 25C Test Level V V IV V IV V IV V V V IV V IV V IV V V V I V I V I V V V I V I V I V V V V V V I V I V I AD9216BCP-65/80 Min Typ Max 58 58 58 TBD AD9216BCP-105 Min Typ Max 57 Unit dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc Bits Bits Bits Bits Bits Bits Bits Bits dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc dBc
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TBD
57 57
57 58 58 58 TBD
56 57
TBD
57 56
56 9.4 9.4 9.4 TBD
55 9.3
TBD
9.3 9.1
9.1 - 70.0 - 69.0 - 70.0
8.9 - 70.0 TBD - 69.0 - 68.0 TBD
- 67.0 - 75.0
- 66.0
- 74.0
TBD
75.0 75.0 75.0 TBD
75.0
74.0 74.0
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AD9216
fINPUT = 100 MHz CROSSTALK
Specifications subject to change without notice.
Preliminary Technical Data
25C Full V V - 80.0 - 80.0 dBc dB
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Preliminary Technical Data ABSOLUTE MAXIMUM RATINGS
Table 5. AD9216 Absolute Maximum Ratings 1
Parameter Pin Name ELECTRICAL AVDD DRVDD AGND AVDD Digital Outputs CLK, DCS, MUX_SELECT, SHARED_REF, OEB, DFS VINA, VINB VREF SENSE REFB, REFT PDWN 2 ENVIRONMENTAL Operating Temperature Junction Temperature Lead Temperature (10 sec) Storage Temperature
1
AD9216
With Respect To AGND DRGND DRGND DRVDD DRGND AGND AGND AGND AGND AGND AGND
Rating Min - 0.3 - 0.3 - 0.3 - 3.9 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 - 45
Max +3.9 +3.9 +0.3 +3.9 DRVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 AVDD + 0.3 +85 +150 +300 +150
Unit V V V V V V V V V V V C C C C
- 65
Absolute maximum ratings are limiting values to be applied individually, and beyond which the serviceability of the circuit may be impaired. Functional operability is not necessarily implied. Exposure to absolute maximum rating conditions for an extended period of time may affect device reliability. 2 Typical thermal impedances (64-lead LQFP); ? ?JA = 54C/W. These measurements were taken on a 4-layer board in still air, in accordance with EIA/JESD51-7.
EXPLANATION OF TEST LEVELS I II 100% production tested. 100% production tested at 25C and sample tested at specified temperatures.
III Sample tested only. IV Parameter is guaranteed by design and characterization testing. V Parameter is a typical value only.
VI 100% production tested at 25C; guaranteed by design and characterization testing for industrial temperature range; 100% production tested at temperature extremes for military devices.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
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AD9216
Table 6. ORDERING GUIDE
Model AD9216BCP-65 AD9216BCPZ-80 AD9216BCPZ-105 AD9216BCPZRL7-65 AD9216BCPZRL7-80 AD9216BCPZRL7105 AD9216-65PCB AD9216-40PCB AD9216-105PCB Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C Package Description
Preliminary Technical Data
Package Option
64-Lead Lead Frame Chip Scale Package (LFCSP) 64-Lead Lead Frame Chip Scale Package (LFCSP) 64-Lead Lead Frame Chip Scale Package (LFCSP) 64-Lead Lead Frame Chip Scale Package (LFCSP) 64-Lead Lead Frame Chip Scale Package (LFCSP) 64-Lead Lead Frame Chip Scale Package (LFCSP) Evaluation Board with AD9216BCPZ-65 Evaluation Board with AD9216BCPZ-80 Evaluation Board with AD9216BCPZ-105
SHARED_REF
MUX_SELECT PDWN_A
57 D9_A (MSB) 56 D8_A 55 D7_A
D6_A DRGND
DRVDD D5_A 52 51
OEB_A OTR_A
AVDD CLK_A
64
63 62
61 60
59 58
54 53
50
AGND VIN+_A VIN-_A AGND AVDD REFT_A REFB_A
1 2 3 4 5 6 7 8
49
D4_A D3_A
48 47 46 45 44 43 42 41 40 39 38 37 36 35 34 33
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
D2_A D1_A D0_A DNC DNC DNC DNC DRVDD DRGND OTR_B D9_B (MSB) D8_B D7_B D6_B D5_B D4_B
VREF SENSE 9 REFB_B 10 REFT_B 11 AVDD AGND VIN-_B 12 13
AD9216 64 Lead for LF-CSP TOP VIEW (Not to Scale)
14 VIN+_B 15 AGND 16
AVDD CLK_B
DCS DFS
PDWN_B OEB_B DNC
DNC D0_B
D1_B D2_B
Figure 3. Pin Configuration
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DRGND DRVDD
D3_B
DNC
DNC
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Preliminary Technical Data
Table 7. Pin Function Descriptions
Pin Number 2 3 15 14 6 7 11 10 8 9 18 63 19 20 21 60 22 59 46-51, 5457 27, 30-38 39 58 62 61 5, 12, 17, 64 1, 4, 13, 16 28, 40, 53 29, 41, 52 23-26, 4245 Mnemonic VIN+_A VIN-_A VIN+_B VIN- _B REFT_A REFB_A REFT_B REFB_B VREF SENSE CLK_B CLK_A DCS DFS PDWN_B PDWN_A OEB_B OEB_A D0_A (LSB)-D9_A (MSB) D0_B (LSB) -D9_B (MSB) OTR_B OTR_A SHARED_REF MUX_SELECT AVDD AGND DRGND DRVDD DNC Description
AD9216
Analog Input Pin (+) for Channel A Analog Input Pin (- ) for Channel A Analog Input Pin (+) for Channel B Analog Input Pin (- ) for Channel B Differential Reference (+) for Channel A Differential Reference (- ) for Channel A Differential Reference (+) for Channel B Differential Reference (- ) for Channel B Voltage Reference Input/Output Reference Mode Selection Clock Input Pin for Channel B Clock Input Pin for Channel A Enable Duty Cycle Stabilizer (DCS) Mode ( Tie to AVDD to enable) Data Output Format Select Bit (Low for Offset Binary, High for Twos Complement) Power-Down Function Selection for Channel B (Active High) Power-Down Function Selection for Channel A (Active High) Output Enable Bit for Channel B (Low Setting Enables Channel B Output Data Bus) Output Enable Bit for Channel A (Low Setting Enables Channel A Output Data Bus) Channel A Data Output Bits Channel B Data Output Bits Out-of-Range Indicator for Channel B Out-of-Range Indicator for Channel A Shared Reference Control Bit (Low for Independent Reference Mode, High for Shared Reference Mode) Data Multiplexed Mode. (See description for how to enable; high setting disables output data Multiplexed mode) Analog Power Supply Analog Ground Digital Output Ground Digital Output Driver Supply. Must be decoupled to DRGND with a minimum 0.1 F capacitor. Recommended decoupling is 0.1 F capacitor in parallel with 10 F Do Not Connect Pins. Should be left floating.
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AD9216
TERMINOLOGY
Aperture Delay Aperture delay is a measure of the sample-and-hold amplifier (SHA) performance and is measured from the rising edge of the clock input to when the input signal is held for conversion. Aperture Jitter The variation in aperture delay for successive samples, which is manifested as noise on the input to the A/D converter. Integral Nonlinearity (INL) INL refers to the deviation of each individual code from a line drawn from negative full scale through positive full scale. The point used as negative full scale occurs 1/2 LSB before the first code transition. Positive full scale is defined as a level 1 1/2 LSB beyond the last code transition. The deviation is measured from the middle of each particular code to the true straight line. Differential Nonlinearity (DNL, No Missing Codes) An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from this ideal value. Guaranteed no missing codes to 10-Bits resolution indicates that all 2048 codes must be present over all operating ranges. Offset Error The major carry transition should occur for an analog value 1/2 LSB below VIN+ = VIN- . Offset error is defined as the deviation of the actual transition from that point. Gain Error The first code transition should occur at an analog value 1/2 LSB above negative full scale. The last transition should occur at an analog value 1 1/2 LSB below the nominal full scale. Gain error is the deviation of the actual difference between first and last code transitions and the ideal difference between first and last code transitions. Temperature Drift The temperature drift for zero error and gain error specifies the maximum change from the initial (25C) value to the value at TMIN or TMAX. Power Supply Rejection The specification shows the maximum change in full scale from the value with the supply at the minimum limit to the value with the supply at its maximum limit. Total Harmonic Distortion (THD) The ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal, expressed as a percentage or in decibels relative to the peak carrier signal (dBc).
Preliminary Technical Data
Signal-to-Noise and Distortion (S/N+D, SINAD) Ratio The ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, including harmonics but excluding dc. The value for S/N+D is expressed in decibels relative to the peak carrier signal (dBc). Effective Number of Bits (ENOB) Using the following formula:
ENOB = (SINAD - 1.76 ) 6.02
effective number of bits for a device for sine wave inputs at a given input frequency can be calculated directly from its measured SINAD. Signal-to-Noise Ratio (SNR) The ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, excluding the first six harmonics and dc. The value for SNR is expressed in decibels relative to the peak carrier signal (dBc). Spurious Free Dynamic Range (SFDR) The difference in dB between the rms amplitude of the input signal and the peak spurious signal. Nyquist Sampling When the frequency components of the analog input are below the Nyquist frequency (fCLOCK/2), this is often referred to as Nyquist sampling. IF Sampling Due to the effects of aliasing, an ADC is not necessarily limited to Nyquist sampling. Higher sampled frequencies will be aliased down into the first Nyquist zone (DC - fCLOCK/2) on the output of the ADC. Care must be taken that the bandwidth of the sampled signal does not overlap Nyquist zones and alias onto itself. Nyquist sampling performance is limited by the bandwidth of the input SHA and clock jitter (jitter adds more noise at higher input frequencies). Two-Tone SFDR The ratio of the rms value of either input tone to the rms value of the peak spurious component. The peak spurious component may or may not be an IMD product. Out-of-Range Recovery Time Out-of-range recovery time is the time it takes for the A/D converter to reacquire the analog input after a transient from 10% above positive full scale to 10% above negative full scale, or from 10% below negative full scale to 10% below positive full scale.
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Preliminary Technical Data
Crosstalk Coupling onto one channel being driven by a (- 0.5 dBFS) signal when the adjacent interfering channel is driven by a full-
AD9216
scale signal. Measurement includes all spurs resulting from both direct coupling and mixing components.
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AD9216
Preliminary Technical Data
TYPICAL PERFORMANCE CHARACTERISTIC PLOTS (TBD)
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Preliminary Technical Data EQUIVALENT CIRCUITS
AD9216
Figure xx. Equivalent Analog Input Circuit
Figure xx. Equivalent Digital Output Circuit
Figure xx. Equivalent Digital Input Circuit
THEORY OF OPERATION
The AD9216 consists of two high performance analog-todigital converters (ADCs) that are based on the AD9215 converter core. The dual ADC paths are independent, except for a shared internal band gap reference source, VREF. Each of the ADC's paths consists of a proprietary front end sample-andhold amplifier (SHA) followed by a pipelined switched capacitor ADC. The pipelined ADC is divided into three sections, consisting of a 4-bit first stage followed by five 1.5bit stages and a final 3-bit fl ash. Each stage provides sufficient overlap to correct for fl ash errors in the preceding stages. The quantized outputs from each stage are combined through the digital correction logic block into a final 10-bit result. The pipelined architecture permits the first stage to operate on a new input sample, while the remaining stages operate on preceding samples. Sampling occurs on the rising edge of the respective clock. Each stage of the pipeline, excluding the last, consists of a low resolution fl ash ADC and a residual multiplier to drive the next stage of the pipeline. The residual multiplier uses the fl ash ADC output to control a switched capacitor digital-to-analog converter (DAC) of the same resolution. The DAC output is subtracted from the stage's input signal and the residual is amplified (multiplied) to drive the next pipeline stage. The
residual multiplier stage is also called a multiplying DAC (MDAC). One bit of redundancy is used in each one of the stages to facilitate digital correction of flash errors. The last stage simply consists of a flash ADC. The input stage contains a differential SHA that can be configured as ac- or dc-coupled in differential or single-ended modes. The output-staging block aligns the data, carries out the error correction, and passes the data to the output buffers. The output buffers are powered from a separate supply, allowing adjustment of the output voltage swing.
ANALOG INPUT
The analog input to the AD9216 is a differential switched capacitor, SHA, that has been designed for optimum performance while processing a differential input signal. The SHA input accepts inputs over a wide common-mode range. An input common-mode voltage of mid supply is recommended to maintain optimal performance. The SHA input is a differential switched capacitor circuit. In Figure 4, the clock signal alternatively switches the SHA between sample mode and hold mode. When the SHA is switched into sample mode, the signal source must be capable of charging the sample capacitors and settling within one-half of a clock cycle. A small resistor in series with each input can
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AD9216
help reduce the peak transient current required from the output stage of the driving source. Also, a small shunt capacitor can be placed across the inputs to provide dynamic charging currents. This passive network will create a low-pass filter at the ADC's input; therefore, the precise values are dependant on the application. In IF under sampling applications, any shunt capacitors should be removed. In combination with the driving source impedance, they would limit the input bandwidth. For best dynamic performance, the source impedances driving VIN+ and VIN- should be matched such that common- mode settling errors are symmetrical. These errors will be reduced by the common-mode rejection of the ADC.
Preliminary Technical Data
levels are defined as follows:
VCM MIN = V REF 2 VCM MAX = (AVDD + V REF
)2
The minimum common- mode input level allows the AD9216 to accommodate ground-referenced inputs. Although optimum performance is achieved with a differential input, a singleended source may be driven into VIN+ or VIN- . In this configuration, one input will accept the signal, while the opposite input should be set to mid-scale by connecting it to an appropriate reference. For example, a 2 VP-P signal may be applied to VIN+ while a 1 V reference is applied to VIN- . The AD9216 will then accept an input signal varying between 2 V and 0 V. In the single-ended configuration, distortion performance may degrade significantly as compared to the differential case. However, the effect will be less noticeable at lower input freque ncies and in the lower speed grade models (AD9216-65 and AD9216-80). Differential Input Configurations As previously detailed, optimum performance will be achieved while driving the AD9216 in a differential input configuration. For base band applications, the AD8138 differential driver provides excellent performance and a flexible interface to the ADC. The output common- mode voltage of the AD8138 is easily set to AVDD/2, and the driver can be configured in a Sallen-Key filter topology to provide band limiting of the input signal. At input frequencies in the second Nyquist zone and above, the performance of most amplifiers will not be adequate to achieve the true performance of the AD9216. This is especially true in IF under sampling applications where freque ncies in the 70 MHz to 200 MHz range are being sampled. For these applications, differential transformer coupling is the recommended input configuration, as shown in Figure 5.
Figure 4. Switched Capacitor Input
An internal differential reference buffer creates positive and negative reference voltages, REFT and REFB, respectively, that define the span of the ADC core. The output common-mode of the reference buffer is set to midsupply, and the REFT and REFB voltages and span are defined as follows:
REFT = 1 2 (AVDD + V REF )
REFB = 1 2 ( AVDD - V REF )
Span = 2 x (REFT - REFB) = 2 x V REF
AD9216
It can be seen from the equations above that the REFT and REFB voltages are symmetrical about the mid-supply voltage and, by definition, the input span is twice the value of the VREF voltage. The internal voltage reference can be pin-strapped to fixed values of 0.5 V or 1.0 V, or adjusted within the same range as discussed in the Internal Reference Connection section. Maximum SNR performance will be achieved with the AD9216 set to the largest input span of 2 VP-P. The relative SNR degradation will be 3 dB when changing from 2 VP-P mode to 1 VP-P mode. The SHA may be driven from a source that keeps the signal peaks within the allowable range for the selected reference voltage. The minimum and maximum common-mode input
Figure 5. Differential Transformer Coupling
The signal characteristics must be considered when selecting a transformer. Most RF transformers will saturate at frequencies below a few MHz, and excessive signal power can also cause core saturation, which leads to distortion. Single-Ended Input Configuration A single-ended input may provide adequate performance in
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Preliminary Technical Data
cost-sensitive applications. In this configuration, there will be a degradation in SFDR and in distortion performance due to the large input common- mode swing. However, if the source impedances on each input are matched, there should be little effect on SNR performance.
AD9216
(by gating, dividing, or other methods), it should be retimed by the original clock at the last step.
POWER DISSIPATION AND STANDBY MODE
The power dissipated by the AD9216 is proportional to its sampling rates. The digital (DRVDD) power dissipation is determined primarily by the strength of the digital drivers and the load on each output bit. The digital drive current can be calculated by
I DRVDD = V DRVDD x CLOAD x fCLOCK x N
CLOCK INPUT AND CONSIDERATIONS
Typical high speed ADCs use both clock edges to generate a variety of internal timing signals, and as a result may be sensitive to clock duty cycle. Commonly, a 5% tolerance is required on the clock duty cycle to maintain dynamic performance characteristics. The AD9216 provides separate clock inputs for each channel. The optimum performance is achieved with the clocks operated at the same frequency and phase. Clocking the channels asynchronously may degrade performance significantly. In some applications, it is desirable to skew the clock timing of adjacent channels. The AD9216's separate clock inputs allow for clock timing skew (typ ically 1 ns) between the channels without significant performance degradation. The AD9216 contains two clock duty cycle stabilizers, one for each converter, that retime the non-sampling edge, providing an internal clock with a nominal 50% duty cycle. Faster Input clock rates (where it becomes difficult to maintain 50% duty cycles) can benefit from using DCS as a wide range of input clock duty cycles can be accommodated. Maintaining a 50% duty cycle clock is particularly important in high speed applicatio ns, when proper track-and-hold times for the converter are required to maintain high performance. The DCS can be enabled by tying the DCS pin high. The duty cycle stabilizer utilizes a delay locked loop to create the non-sampling edge. As a result, any changes to the sampling frequency will require approximately 2 s to 3 s to allow the DLL to acquire and settle to the new rate. High speed, high resolution ADCs are sensitive to the quality of the clock input. The degradation in SNR at a given full-scale input frequency (fINPUT ) due only to aperture jitter (tJ ) can be calculated with the following equation:
SNR deg radation = 20 x log 10 1 2 x p x f INPUT x t J
where N is the number of bits changing and CLOAD is the average load on the digital pins that changed. The analog circuitry is optimally biased so that each speed grade provides excellent performance while affording reduced power consumption. Each speed grade dissipates a baseline power at low sample rates that increases with clock frequency. Either channel of the AD9216 can be placed into standby mode independently by asserting the PWDN_A or PDWN_B pins. It is recommended that the input clock(s) and analog input(s) remain static during either independent or total standby, which will result in a typical power consumption of 1 mW for the ADC. Note that if DCS is enabled, it is mandatory to disable the clock of an independently powered-down channel. Otherwise, significant distortion will result on the active channel. If the clock inputs remain active while in total standby mode, typical power dissipation of TBD mW will result. The minimum standby power is achieved when both channels are placed into full power-down mode (PDWN_A = PDWN_B = HI). Under this condition, the internal references are powered down. When either or both of the channel paths are enabled after a power-down, the wake-up time will be directly related to the recharging of the REFT and REFB decoupling capacitors and to the duration of the power-down. Typically, it takes approximately 5 ms to restore full operation with fully discharged 0.1 F and 10 F decoupling capacitors on REFT and REFB.
[
]
In the equation, the rms aperture jitter, tJ , represents the rootsum square of all jitter sources, which includes the clock input, analog input signal, and ADC aperture jitter specification. Under-sampling applications are particularly sensitive to jitter. For optimal performance, especially in cases where aperture jitter may affect the dynamic range of the AD9216, it is important to minimize input clock jitter. The clock input circuitry should use stable references, for example using analog power and ground planes to generate the valid high and low digital levels for the AD9216 clock input. Power supplies for clock drivers should be separated from the ADC output driver supplies to avoid modulating the clock signal with digital noise. Low jitter crystal controlled oscillators make the best clock sources. If the clock is generated from another type of source
A single channel can be powered down for moderate power savings. The powered-down channel shuts down internal circuits, but both the reference buffers and shared reference remain powered. Because the buffer and voltage reference remain powered, the wake-up time is reduced to several clock cycles.
DIGITAL OUTPUTS
The AD9216 output drivers can be configured to interface with 2.5 V or 3.3 V logic families by matching DRVDD to the digital supply of the interfaced logic. The output drivers are sized to provide sufficient output current to drive a wide variety of logic families. However, large drive currents tend to cause current glitches on the supplies that may affect converter performance. Applications requiring the ADC to drive large capacitive loads or large fan-outs may require external buffers or latches.
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AD9216
The data format can be selected for either offset binary or twos complement. This is discussed later in the Data Format section.
Preliminary Technical Data
AD9216 using the DCS pin. This provides a stable 50% duty cycle to internal circuits. The length of the output data lines and loads placed on them should be minimized to reduce transients within the AD9216. These transients can detract from the converter's dynamic performance. The lowest typical conversion rate of the AD9216 is 1 MSPS. At clock rates below 1 MSPS, dynamic performance may degrade.
TIMING
The AD9216 provides latched data outputs with a pipeline delay of six clock cycles. Data outputs are available one propagation delay (tPD ) after the rising edge of the clock signal. Refer to Figure 2 for a detailed timing diagram. The internal duty cycle stabilizer can be enabled on the
Figure 6. NEEDS UPDATING Example of Multiplexed Data Format Using the Channel A Output and the Same Clock Tied to CLK_A, CLK_B, and MUX_SELECT
DATA FORMAT
The AD9216 data output format can be configured for either twos complement or offset binary. This is controlled by the Data Format Select pin (DFS). Connecting DFS to AGND will produce offset binary output data. Conversely, connecting DFS to AVDD will format the output data as twos complement. The output data from the dual A/D converters can be multiplexed onto a single 10-Bits output bus. The multiplexing is accomplished by toggling the MUX_SELECT bit, which directs channel data to the same or opposite channel data port. When MUX_SELECT is logic high, the Channel A data is directed to Channel A output bus, and Channel B data is directed to the Channel B output bus. When MUX_SELECT is logic low, the channel data is reversed, i.e., Channel A data is directed to the Channel B output bus and Channel B data is directed to the Channel A output bus. By toggling the MUX_SELECT bit, multiplexed data is available on either of the output data ports. If the ADCs are run with synchronized timing, this same clock can be applied to the MUX_SELECT bit. After the MUX_SELECT rising edge, either data port will ha ve the data for its respective channel; after the falling edge, the alternate channel's data will be placed on the bus. Typically, the other unused bus would be disabled by setting the appropriate OEB high to reduce power consumption and noise. Figure 6 shows an example of multiplex mode. When multiplexing data, the data rate is two times the sample rate. Note that both channels
must remain active in this mode and that each channel's powerdown pin must remain low.
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Preliminary Technical Data
VOLTAGE REFERENCE
A stable and accurate 0.5 V voltage reference is built into the AD9216. The input range can be adjusted by varying the reference voltage applied to the AD9216, using either the internal reference with different external resistor configurations or an externally applied reference voltage. The input span of the ADC tracks reference voltage changes linearly. If the ADC is being driven differentially through a transformer, the reference voltage can be used to bias the center tap (common mode voltage). The Shared Reference mode allows the user to connect the references from the dual ADCs together externally for superior gain and offset matching performance. If the ADCs are to function independently, the reference decoupling can be treated independently and can provide superior isolation between the dual channels. To enable Shared Reference mode, the SHARED_REF pin must be tied high and external differential references must be externally shorted. (REFT_A must be externally shorted to REFT_B and REFB_A must be shorted to REFB_B.) Internal Reference Connection A comparator within the AD9216 detects the potential at the SENSE pin and configures the reference into four possible states, which are summarized in Table 8. If SENSE is grounded, the reference amplifier switch is connected to the internal resistor divider (see Figure 7), setting V to 1 V. REF Connecting the SENSE pin to VREF switches the reference amplifier output to the SENSE pin, Table 8. Reference Configuration Summary
Selected Mode External Reference Internal Fixed Reference Programmable Reference Internal Fixed Reference SENSE Voltage AVDD VREF 0.2 V to VREF AGND to 0.2 V Resulting VREF (V) N/A 0.5 0.5 x (1 + R2/R1) 1.0
AD9216
completing the loop and provid ing a 0.5 V reference output. If a resistor divider is connected as shown in Figure xx, the switch will again be set to the SENSE pin. This will put the reference amplifier in a non- inverting mode with the VREF output defined as fo llows:
V REF = 0.5 x (1 + R 2 R1)
In all reference configurations, REFT and REFB drive the ADC core and establish its input span. The input range of the ADC always equals twice the voltage at the reference pin for either an internal or an external reference.
AD9216
Figure 7. Internal Reference Configuration
Resulting Differential Span (VP-P) 2 x External Reference 1.0 2 x VREF (See Figure xx) 2.0
External Reference Operation The use of an external reference may be necessary to enhance the gain accuracy of the ADC or to improve thermal drift characteristics. When multiple ADCs track one another, a single reference (internal or external) may be necessary to reduce gain matching errors to an acceptable level. A high precision external reference may also be selected to provide lower gain and offset temperature drift. Figure 10 shows the typical drift characteristics of the internal reference in both 1 V and 0.5 V modes. When the SENSE pin is tied to AVDD, the internal reference will be disabled, allowing the use of an external reference. An internal reference buffer will load the external reference with an equivalent 7 k load. The internal buffer will still generate the positive and negative full-scale references, REFT and REFB, for the ADC core. The input span will always be twice the value of the reference voltage; therefore, the external reference must be limited to a maximum
of 1 V. If the internal reference of the AD9216 is used to drive multiple converters to improve gain matching, the loading of the reference by the other converters must be considered. Figure X depicts how the internal reference voltage is affected by loading.
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AD9216
Preliminary Technical Data
AD9216
Figure xx. VREF Accuracy vs. Load Figure xx. Programmable Reference Configuration
Figure xx. Typical VREF Drift
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Preliminary Technical Data EVALUATION BOARD DIAGRAMS (TBD)
AD9216
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AD9216 OUTLINE DIMENSIONS
Preliminary Technical Data
Figure 8. 64-Lead Lead Frame Chip Scale Package (LFCSP)
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PR04775-0-6/04(PrD)

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